Display driver apparatus

ABSTRACT

An emitter follower amplifier is coupled between the output of a high voltage video amplifier and the cathode of a kinescope for reducing the effective capacitance presented to the video amplifier that is attributable to the kinescope cathode, socket, spark gaps, and related stray capacitances. A secondary undesired capacitance loading of the video amplifier is effectively reduced by regulating the collector-emitter voltages of the emitter follower output transistors at substantially constant values thereby improving parameters such as slew rate and bandwidth of the overall video display system. Nonlinear circuitry in the follower circuit facilitates accurate AKB current sensing and provides simplification of the voltage regulation circuitry. White limiting circuitry is included within the video amplifier for reducing excess peak white drive that results in objectionable smears. Black limiting circuitry within the video amplifier is also provided to prevent the collapse of the collector-emitter voltage of the push-pull follower output transistor. The heat sink capacitance of the video amplifier output transistor is screened out via bootstrapping.

This invention relates to amplifiers generally and particularly to anapparatus for providing amplification of a video signal for driving thecathode electrode of a kinescope.

In television apparatus employing direct view or projection kinescopesas display devices, it is desirable that the amplifier driving thekinescope cathode provide a relatively high voltage drive signal havinga wide bandwidth and a high slew rate. Typically, drive voltages may beon the order of 200 volts or more and bandwidths may be substantiallyhigher than conventional television standards in certain applications,for example, where both conventional TV viewing and data display aredesired. Even higher bandwidths may be required in video applicationsrequiring scanning at two times of more of the standard TV line rate.

To facilitate high voltage operation it is common to employ a cascodeconfiguration of a common emitter input stage driving a common baseoutput stage. Such a configuration requires only one high voltagetransistor (the output stage) and since it is connected in a common baseconfiguration the Miller effect is suppressed and very wide bandwidthoperation is thus possible. In practice, the actual bandwidth and slewrate which may be achieved in a cascode amplifier depends, to a greatextent, on the effective load capacitance presented to the output stageand the available output current.

In general, one may either increase the amplifier operating current ordecrease the effective load capacitance to maximize the bandwidth andslew rate of the amplifier. However, since increasing the currentnecessarily implies increasing the amplifier power dissipation, it ispreferable to take steps to reduce the effective load capacitance forimproved performance rather than resort to increases in operating power.

In kinescope driver applications, the “effective” load capacitancepresented to the amplifier is principally that of the kinescope cathodeand stray capacitances associated with the socket, spark gaps, wiringand the like. An effective approach to reducing the effectivecapacitance loading is to couple the amplifier to the cathode by meansof a push-pull complementary emitter follower amplifier. Such anamplifier effectively “isolates” the load capacitance approximately inproportion to the reciprocal of the transistor current gain (“beta”).The additional current provided by the follower amplifier providesfaster charging and discharging of the load capacitance and thusenhances slew rate and bandwidth. To avoid substantially increasing thequiescent power dissipation, it is customary practice to operate thefollower amplifier in a “class-B” mode in which the push-pulltransistors are biased to avoid simultaneous conduction.

An example of a kinescope driver amplifier in which the load of acascode amplifier is coupled to the cathode of a kinescope via apush-pull complementary emitter follower amplifier for load capacitancereduction is described, for example, by John H. Furrey, in U.S. Pat. No.4,860,107 entitled VIDEO DISPLAY DRIVER APPARATUS which issued Aug. 22,1989. Advantageously, the use of a complementary emitter follower driverin the Furrey apparatus significantly reduces the effective loadcapacitance of the display (kinescope load and stray capacitances)thereby improving positive and negative video signal transient response.

It has been recognized by White et al. in U.S. Pat. No. 5,680,173entitled KINESCOPE DRIVER APPARATUS, which issued Oct. 21, 1997, thatsignificant further improvements may be made in kinescope driveramplifiers of a type having a complementary emitter follower outputcoupling stage. Specifically, in the White et al. apparatus a push-pullcomplementary emitter follower amplifier is coupled between the outputof a high voltage driver amplifier and the cathode of a kinescope forreducing the effective capacitance presented to the driver amplifierthat is attributable to the kinescope cathode, socket, spark gaps andrelated stray capacitance. A secondary undesired capacitance loading ofthe amplifier attributable to the collector-to-base capacitances of thefollower amplifier is effectively reduced by regulating thecollector-to-emitter voltages of the push-pull follower outputtransistors at respective substantially constant values therebyimproving parameters such as the slew rate and bandwidth of the overallvideo display system.

In the prior art discussed above, a desirable reduction in loadcapacitance has been achieved by using emitter follower load isolation(Furrey) and a further reduction in load capacitance was achieved byregulating the collector-emitter voltages of the follower transistors(White et al.).

It is herein recognized that a need exists for further improvements inkinescope driver amplifiers of a type employing positive feedback foremitter follower capacitance reduction in applications where it is alsodesired to provide accurate sensing of the kinescope cathode current forpurposes such as automatic kinescope bias (hereafter “AKB”) control. Thepresent invention is directed, in a first respect, to meeting that need.

The present invention relates to kinescope driver apparatus of a typecomprising a complementary emitter-follower amplifier having an inputcoupled to a video amplifier for receiving a video signal and having anoutput coupled to a cathode of a kinescope. A feedback circuit isprovided for applying respective positive feedback voltages torespective collectors of first and second output transistors of thecomplementary emitter follower for maintaining substantially constantcollector-to-emitter voltages for the output transistors and an AKBcurrent sensor is connected in a collector circuit of the second outputtransistor for sensing cathode current of the kinescope.

In accordance with the invention, the output of the complementaryemitter follower is coupled via a threshold conduction switch means tothe emitter of the first output transistor and is coupled via resistivemeans to the emitter of the second output transistor.

In accordance with a further feature of the invention, the feedbackcircuit has an input connected to a junction of the threshold conductionswitch means and the emitter of the first output transistor.

In accordance with another feature of the invention, a capacitor iscoupled in parallel with the threshold conduction switch means betweenthe emitter of the first output transistor and the output of thecomplementary emitter follower.

The foregoing and further features of the invention are illustrated inthe accompanying drawings, wherein like elements are designated by likereference numbers, and in which:

FIG. 1 is a schematic diagram, partially in block form, of a prior artkinescope driver apparatus having complementary emitter follower cathodeisolation and AKB current sensing;

FIG. 2 is a schematic diagram, partially in block form, of anotherembodiment of a prior art kinescope driver apparatus;

FIG. 3 is a schematic diagram, partially in block form, of a kinescopedriver apparatus embodying the invention; and

FIGS. 4 and 5 are block diagrams, partially in schematic form,illustrating further features of the invention for effecting loadcapacitance reduction in conjunction with the apparatus of FIG. 3.

It is helpful to an understanding of the present invention to firstconsider the FIG. 1 which is an embodiment of the kinescope driverapparatus of White et al. in U.S. Pat. No. 5,680,173 mentioned above,and the problem of using conventional push-pull emitter follower bufferamplifiers for isolating the kinescope cathode capacitance from theoutput of the kinescope driver amplifier. As previously explained, theemitter follower amplifier is effective in reducing the capacitanceattributable to the cathode (and associated strays) that is presented tothe output of the high voltage video driver amplifier. However, thefollower amplifier itself may introduce a capacitance loading effect onthe driver amplifier and may tend to limit the overall systemperformance.

White et al. point out that the main source of undesirable capacitanceloading effects in kinescope driver systems of the type using push-pullemitter followers is attributable to the collector to base capacitancesof the follower output transistors. Typically, these capacitances aresmaller than the kinescope cathode capacitance and isolating the cathodeby an emitter follower amplifier provides an overall capacitancereduction and improvement in slew rate and bandwidth as compared withdirect coupled systems. However, to achieve the maximum benefit ofemitter follower isolation, it is desirable to reduce the effectivecapacitance of the emitter follower amplifier itself.

To achieve an effective reduction in follower capacitance in the Whiteet al. apparatus, feedback is used in such a manner as to reduce theflow of current under dynamic signal conditions in the collector to basecapacitances of the follower transistors. This is achieved by applyingthe feedback to maintain a substantially constant collector to emittervoltage for the follower transistors. This maintains a constantcollector to base voltage. As a result, under dynamic signal conditions,there is little or no charging or discharging of the collector to basecapacitances as the signal voltage varies.

The effective reduction in follower input capacitance attributable tothe transistor collector to base capacitances is a function of thepercentage of feedback applied to regulating the collector to emittervoltage. If, for example, the feedback percentage is selected such thatcollector to emitter voltage variations are reduced by fifty percent,then the reactive currents charging and discharging the collector tobase capacitances of the follower amplifier will be also reduced byfifty percent and the “effective” capacitance loading will be cut inhalf. Greater reductions in follower capacitance may be achieved as thefeedback percentage is increased towards unity. For purposes of circuitstability, provisions were made to ensure that the feedback gain cannotequal or exceed unity. This is achieved by connecting all “active”semiconductor devices in the feedback paths in voltage or “emitter”follower configurations.

FIG. 1 herein illustrates an embodiment of the White et al. apparatuswhich includes cathode current sensing for AKB control and is describedherein in order to provide a foundation for the improvements of thepresent invention. FIG. 1 shows a television display system including avideo signal source 10 for supplying a video signal to a kinescopecathode 16 for display. To simplify the drawing, the details of thekinescope and the signal source are not shown. It will be appreciatedthat for a color system there would be three driver amplifiers.

As an overview, to amplify the video signal to the high voltage levelsrequired at cathode. 16 the system includes a cascode type of highvoltage amplifier 20 (outlined in phantom). To isolate the output of thehigh voltage amplifier 20 from the capacitance of the kinescope cathode16 the output of amplifier 20 (collector of transistor Q3) is coupled tocathode 16 via a push pull complementary emitter follower amplifier 30(outlined in phantom). To protect the driver amplifier from kinescopearcs, the follower output terminal 15 is coupled to the cathode 16 bymeans of a kinescope arc protection resistor R15 and inductor L1. Toprovide for automatic kinescope bias (AKB) operation, a cathode currentsensing circuit 40 (“I^(k) sense”, outlined in phantom) is providedwhich senses the collector current of a PNP transistor (Q7) in thepush-pull emitter follower amplifier 30 to generate an AKB output signalat output terminal 18 proportional to the cathode current, I_(k), of thekinescope cathode 16. This feature is optional and may be omitted.

Finally, to reduce the effective capacitance presented to the highvoltage amplifier which is attributable to the collector to basecapacitances of the complementary emitter follower 30, the systemincludes a feedback control circuit 50 (outlined in phantom) whichmaintains a substantially constant collector emitter voltage for the NPNtransistor Q4 of the follower 30 and another feedback control circuit 60(outlined in phantom) which maintains a substantially constant collectoremitter voltage for the PNP transistor Q7 of follower 30. The operationof the follower transistors at constant collector to emitter voltagesmaintains the collector to base voltages at a nearly constant valuereducing the magnitude of the charging and discharging currents of thecollector to base capacitances of the follower transistors. Thebeneficial result is that, since the driver amplifier 20 does not haveto supply charging and discharging currents for these “parasitic”capacitances, the overall slew rate, bandwidth and transient responsecharacteristics are improved.

High voltage power (e.g., 200 volts or so) for operation of theamplifier 20 and the feedback or regulator circuits 50 and 60 isprovided by high voltage (H.V.) supply terminal 20. Decoupling of thehigh voltage supply (20) is provided by a decoupling network or low passfilter comprising resistor R20 and capacitor C20. A low voltage (L.V.)supply terminal 21 provides a relatively low voltage (e.g., 12 volts orso) for biasing the input and cascode stages of the high voltage videodriver amplifier 20. This supply input is also decoupled by means of anRC network comprising resistor R21 and capacitor C21.

The high voltage driver amplifier 20 comprises an NPN common emitterconnected input transistor Q2 connected in cascode with a common baseconnected NPN output transistor Q3. A fixed base bias voltage for thecascode output transistor Q3 is provided by the low voltage (e.g., +12volts) decoupling network (R21, C21). A lower potential for operation ofthe emitter load resistor R6 of the input transistor Q2 is provided by aZener diode regulator comprising resistor R5 and Zener diode CR1 coupledbetween the base of transistor Q3 and ground. Illustratively, the Zenervoltage may be 5 or 6 volts which establishes a DC reference for theload resistor R6 of the cascode input transistor as well as a DCreference for the AKB sense amplifier 40. The emitter electrode of theinput transistor Q2 is also coupled to ground via a high frequencypeaking network comprising resistor R7 and capacitor C2 which arecoupled in series.

The video input signal to be amplified, provided by source 10, isapplied to the base of the cascode input transistor via an emitterfollower input stage comprising PNP transistor Q1 which is connected atthe collector thereof to ground and coupled at the base thereof to thevideo input terminal 12 via an input resistor R3. The emitter oftransistor Q1 is coupled to the base of transistor Q2 and to the lowvoltage supply 21 via an emitter resistor R4. Additional high frequencypeaking is provided by a further peaking network comprising seriesconnected resistor R1 and capacitor C1 coupled in parallel with theinput resistor R3.

The collector load for the cascode amplifier 20 is provided by resistorR8 which is coupled from the high voltage supply 10 to the collector ofthe cascode output transistor Q3. A diode CR2 is interposed between theload resistor R8 and the collector of transistor Q3 to provide a smalloffset voltage for reducing cross-over distortion in the complementaryemitter follower amplifier 30.

During operation of the cascode amplifier 20, the open loop gain isdirectly proportional to the value of the load resistor R8 and inverselyproportional to the impedance of the emitter network R6, C2 and R7 aspreviously discussed. The open loop gain, bandwidth and slew rate isalso a function of the capacitive loading of the output of amplifier 20(i.e., the capacitance presented to the collector of transistor Q3).This capacitance is reduced by operating the push-pull transistors ofthe complementary emitter follower amplifier 30 at constant collector toemitter voltages. The closed loop gain, assuming that the open loop gainis adequate, is directly proportional to the value of the feedbackresistor R2 and inversely proportional to the impedance of the inputnetwork R1, R3 and C1.

The push-pull complementary emitter follower amplifier 30 includes apair of complementary transistors Q4 and Q7 coupled at base electrodesthereof to the output (collector of Q3) of amplifier 20 and coupled atthe emitters thereof to an output terminal 15 via respective emitterresistors R9 and R12. The output 15 of emitter follower amplifier 30 iscoupled, as previously noted, to the cathode 16 via a kinescope arcsuppression network comprising the series connection of inductor L1 andresistor R15. Supply voltage (collector potentials) for the followertransistors Q4 and Q7 are provided by respective feedback circuits 50and 60.

Circuit 50 regulates the collector to emitter voltage of followertransistor Q4 at a fixed value and includes a voltage regulatortransistor Q6 connected at the collector thereof to supply 20 and at theemitter thereof to the collector of transistor Q4. The input (base) ofthe voltage regulator transistor Q6 is coupled to the emitter electrodeof the follower transistor Q4 via a capacitor C3 in parallel with athreshold conduction device, Zener diode CR3. This positive feedbackpath establishes a substantially constant collector to emitter offsetvoltage for follower transistor Q4 that is equal to the Zener voltage.To provide an operating current for the Zener diode, the cathode iscoupled to the high voltage source 20 via resistor R11. To minimizeloading of the emitter circuit of transistor Q4, the emitter is coupledto the capacitor C3 and Zener diode CR3 via an emitter followertransistor Q5. Specifically, transistor Q5 is a PNP transistor coupledat its base to the emitter of the follower transistor Q4 via a resistorR10. The collector-emitter path of follower transistor Q5 is coupledbetween the junction of capacitor C3 and Zener diode CR3 and ground.

Circuit 60 is similar to circuit 50 and regulates the collector toemitter voltage of follower transistor Q7 at a fixed value. Circuit 60includes a voltage regulator transistor Q9 connected at its collector toa supply input of the I_(k) sense amplifier 40 and at its emitter to thecollector of transistor Q7. The input of the voltage regulatortransistor Q9 is coupled to the emitter electrode of the followertransistor Q7 via a capacitor C4 in parallel with a threshold conductiondevice, Zener diode CR4. This feedback path regulates the collectoremitter voltage of the follower transistor Q7 to the Zener voltage. Toprovide an operating current for the Zener diode, the anode thereof iscoupled to ground via a resistor R14. To minimize loading of the emittercircuit of transistor Q7, the emitter is coupled to the capacitor C4 andZener diode CR4 via an emitter follower transistor Q8. Specifically,transistor Q8 is a NPN transistor coupled at its base to the emitter ofthe follower transistor Q7 via a resistor R13. The collector-emitterpath of transistor Q8 is coupled between the junction of capacitor C4and Zener diode CR4 and the high voltage supply 20.

The I_(k) sense amplifier 40 is provided for use in video displaysystems of the type featuring automatic kinescope bias (AKB) circuitryand thus requiring sensing of the kinescope cathode current “I_(k)”.Sense amplifier 40 comprises a cathode current sensing transistor Q10connected at the emitter thereof to the collector of the voltageregulator transistor Q9. A reference potential for the base oftransistor Q10 is provided by the Zener diode CR1. Capacitor C5, inparallel with diode CR1 provides filtering of the regulated Zenervoltage. An output voltage, proportional to the cathode current I_(k) isdeveloped at output terminal 18 across the load resistor R16 coupledbetween the collector of transistor Q10 and ground. In applications notrequiring AKB operation the sense amplifier may be omitted. If omittedthen the collector of voltage regulator transistor Q9 should be coupledto ground or another suitable low voltage reference potential.

To summarize the operation described above, the cascode amplifier 20amplifies the video signal provided by source 10 as previouslydescribed. To minimize the capacitive loading on load resistor R8 thatis attributable to the capacitance associated with the kinescope 16, itssocket and spark arrestors (not shown) and other stray capacitances, theoutput (collector of transistor Q3) of the cascode amplifier 20 iscoupled to the kinescope cathode electrode via a push-pull complementaryemitter follower amplifier 30. This particular follower amplifier is ofthe “parallel” type in which the base electrodes are in parallel forreceiving the amplified video signal and the emitters are in parallelfor driving the cathode.

The inclusion of the emitter follower amplifier 30 provides a reductionin cathode capacitance presented to the amplifier 20 but introduces asecondary capacitance effect. Namely, the collector to base capacitancesof follower transistors Q4 and Q7. To effectively reduce the values ofthese unwanted capacitances, the reactive charging and dischargingcurrents supplied to these capacitances are reduced. This feature isprovided by the two positive feedback regulators 50 and 60 whichmaintain the collector to emitter voltages for the follower transistorsat constant values.

As an example, if the output voltage of amplifier 20 increases, then theemitter voltage of follower transistor Q4 will increase but Zener diodeCR3 and regulator transistor Q6 will increase the collector voltage offollower transistor Q4. Similarly, for a decreasing output voltage ofamplifier 20, the emitter voltage of follower transistor Q4 willdecrease and Zener diode CR3 and regulator transistor Q6 will cause adecrease in the collector voltage of follower transistor Q4.Illustratively, for a Zener voltage of 10 Volts, the collector emittervoltage of transistor Q4 will equal the Zener voltage. For the assumedZener voltage of 10 volts, the resultant collector-emitter voltage oftransistor Q4 will be approximately 10 Volts.

Thus, whether the follower input voltage is increasing or decreasing,the voltage across the follower transistor from the collector to theemitter is constant. As the input signal goes through points ofinflection, the base voltage will vary by a few hundred millivoltsrelative to the emitter as the follower transistor is biased on and off(push-pull operation). However, it has been found that base emittervoltage variations are relatively minor as compared with the regulatedcollector emitter voltage (e.g., a Zener voltage of 10 volts or so). Asa result, one may consider collector to base voltage variations“substantially” constant and there can be little charging anddischarging of the collector to base capacitance under dynamic signalconditions. Since such reactive currents are suppressed, the effectivecollector to base capacitances are reduced for the follower amplifier.

As described above, the feedback for regulating the collector emittervoltages for the follower transistors is nearly one hundred percent butcannot equal . unity since that would require infinite current gains oftransistors Q5 and Q6. In other words, transistors Q5 and Q6 are bothconnected as emitter followers and the gain is close to but less thanunity. Accordingly, even though the feedback is positive, the circuit isstable. Lesser amounts of feedback, e.g. 50%, may be used if desired ina given application. It should be noted that the actual Zener voltage isnot a critical parameter of the circuit. The Zener by-pass capacitor (C3or C4) provides a desirable reduction in AC impedance of the voltageregulator to further facilitate wideband operation.

FIG. 2 illustrates a second embodiment of a kinescope driver disclosedby White et al. In this embodiment, a reduction in the overall number ofparts is realized by eliminating transistors Q5 and Q6 and resistors RIOand R13 and connecting feedback control circuits 50B and 60B andfollower amplifier 30A as shown. However, this embodiment is not fullysuitable for use with the AKB sensing described above because adding anAKB sensing circuit to the second embodiment causes unwanted currents toflow through the AKB sensing circuit during AKB intervals therebyadversely affecting the accuracy of AKB control. This effect, amongothers, is addressed in the present invention as discussed below.

FIG. 3 illustrates improvements, in accordance with the presentinvention, over the apparatus of White et al. described above. Thekinescope driver of the present invention also employs emitter followercoupling of a video amplifier to a kinescope with positive feedback forreducing the emitter follower capacitance and with AKB current sensing.Where the output stage emitter follower amplifier of the White apparatusrequired six active components (Q4-Q9), the present invention requiresonly four active components (Q4, Q6, Q7 and Q9). Further improvements,in accordance with the invention, include improved AKB sense operation,improved AC operation of the output stage, the addition of adifferential reference input stage, the addition of white and blacklimiting circuitry in the video amplifier stage, and heat sinkbootstrapping.

As shown in FIG. 3, the AKB sense circuit of the present inventioncomprises an emitter follower transistor Q10 connected at its base inputto the low voltage supply terminal 21, at its collector to a resistordivider network (R16A and R16B), and at its emitter to the collector oftransistor Q9. A capacitor C5 is connected in parallel across thebase/emitter junction of Q10 to provide buffering action.

Improved AKB sensing operation is achieved for DC output voltagesgreater than approximately VCC2(R14/(R14+R11)). The net current intothreshold conduction devices (Zener diodes) CR3 and CR4 is positive andsupplied by transistors Q4 and Q6. For the case when resistor R14 isequal to resistor R11, positive net current into diodes CR3 and CR4occurs for DC output voltages greater than approximately one half thatof the high voltage supply at terminal 20 (e.g., VCC2). This gives morethan enough range for AKB sense cutoff measurements. For DC conditions,the output bias diode network of CR2A and CR2B results in diode CR2Chaving approximately zero volts across it, thus CR2C does not conduct.Under these conditions, with diode CR2C biased off, the DC cathodecurrent has to flow through transistor Q7's emitter electrode and hencethe collector current of transistor Q9 represents the CRT cathodecurrent through inductor L1 and resistor R16 with an error equal to thesum of the reciprocals of the betas of transistors Q7 and Q9.

The buffering action of AKB sense transistor Q10 and capacitor C5provide DC and AC low impedance at the collector of transistor Q9 andalso the necessary limiting. of the “I_(k) Sense” voltage which isproportional to the CRT cathode current. The low impedance at thecollector of transistor Q9 is desirable for maintaining the frequencyresponse of the CRT driver stage. The limiting action is desirablebecause peak cathode currents can reach the 10's of mA, while the AKBcutoff currents are in the 10's of μA. For higher cathode currents,transistor Q10 saturates and its collector voltage is limited toVCC1+Vbe (the base/emitter voltage of transistor Q10). Resistor dividernetwork R16A and R16B further attenuate the peak I_(k) sense voltage. Atcutoff, transistor Q10 operates in its linear region as a common basestage and the voltage at I_(k) is substantially equal to the CRT cathodecurrent multiplied by resistor R16B (assuming high impedance sensing).

The AC Operation of the Boot-Strap output stage is essentially the sameas in the White et al. apparatus. That is, the collector to base inputcapacitances, C_(cb), of transistors Q4 and Q7 are canceled by the nearunity positive voltage feedback to the collectors of said devices forfrequencies appreciably less than F_(T) of Q4 and Q7. Advantageously,the near unity feedback is achieved using one less active component(transistor) in each feedback circuit than required in the White et al.apparatus. Also, capacitor C200 has been provided across thresholdconduction device (diode) CR2C to reduce the small signal AC coring ofthe signal that drives the CRT cathode.

The addition of a reference input circuit (206), an emitter followertransistor (Q1B) stage, causes the collector current through transistorsQ2 and Q3 to be proportional to the voltage difference between the“Video IN” and “Ref IN” inputs (terminals 12A and 12B, respectively),thereby providing good ground difference rejection between the smallsignal sections of the TV or display and the large signal CRT driverstage. (For this purpose, the video signal source 10 provides the videosignal S1 to input 12A and also provides a video signal referencevoltage S2 to input 12B of the cascode amplifier 20.) Lack of adequateground signal rejection could lead to “regeneration”, ringing, andextraneous noise and artifact pickup. By making both the “Video IN”(12A) and “Ref IN” (12B) inputs high impedances, signal radiation fromsignal or ground currents is reduced.

White limiting circuit 200 (outlined in phantom) comprises transistorQ1, diode D1, and resistors R20 and R21. This is desirable becausesaturation of transistors Q3, Q4 or Q7 caused by excess peak white drivecan result in the stretching of momentary overdrive into objectionablesmears. The action of transistor Q1, and resistors R20 and R21 issufficient to provide limiting but the addition of diode D1 softens andproduces a more pleasing limiting action. Additionally, diode D1 resultsin an approximately net zero V_(be) temperature compensation for thedifferential input (Ref IN) 12B.

It may be appreciated that the side of resistor R21 shown attached toground may be attached to the emitter of the reference input transistorQ2 instead of ground. This will provide essentially the same limitingaction but referenced to the “Ref IN” signal rather than to ground.

Although less severe in effect than excess peak whites, excess peak“blacker than black” peaks can result in the collapse of the collectorto emitter voltage, V_(ce), across transistor Q3 and cause anundesirable stretching of these excessively large black transients intowider, more visible artifacts. This condition is eliminated by theaddition of resistor R202 between the emitter of transistor Q3 andground. The DC current flowing through resistor R202 from the commonbase transistor Q3 is selected to prevent collapse of the voltage acrosstransistor Q4 even when there is no current flowing through the emitterof transistor Q4.

The frequency response of the CRT driver and its limiting slew rate aredetermined primarily by the net capacitance (C_(c)) at the collector oftransistor Q3 (the output of the video amplifier) and the value ofresistor R8. Capacitor C2 is chosen so that the product of (R8)(C_(c))is equal to the product of (R7)(C2). This compensates for the smallsignal roll off caused by the net capacitance, C_(c), of transistor Q3and resistor R8. During large black-going transitions, however, thiscompensation does not work since the collector current of transistors Q3and Q2 cannot go negative.

It is desirable to reduce the effective value of C_(c) as much aspossible in order to produce the best large signal response for a givenvalue of resistor R8 which sets the power dissipation of transistor Q3.The sources of net capacitance, C_(c), of transistor Q3 include theinput capacitances of transistors Q4 and Q7, the collector-basecapacitance, C_(cb), of transistor Q3, the wiring capacitance, and thecapacitance of the heat sink for transistor Q3.

By bootstrapping the collector electrodes of transistors Q4 and Q7 andoperating them as emitter followers, the input capacitances oftransistors Q4 and Q7 are virtually eliminated.

The heat sink capacitance of transistor Q3 shows up as a capacitor fromthe collector of transistor Q3 to the actual heat sink which istypically a metal assembly. The capacitance added by the heat sink oftransistor Q3 can be screened out or “bootstrapped” by electricallyconnecting the heat sink of transistor Q3 either to the signal at outputterminal 15 or at the emitter of transistors Q4 or Q7. The voltage atthe emitters of transistors Q4 and Q7 follows the voltage at thecollector of transistor Q3 and has a positive gain that is slightly lessthan one.

FIG. 4 shows apparatus for reducing the effective capacitance of theheat sink by positive feedback. Here, the output at the emitters oftransistors Q4 or Q7 or at the output terminal 15 is applied either byDC or AC coupling to the heat sink 500 of transistor Q3. Transistor Q3is thermally coupled to the heat sink 500. For DC shielding, the outputof transistors Q4 or Q7 or the output terminal 15 is directly applied toheat sink 500 via terminal 502 or is AC coupled via capacitor 506 and ACcoupling terminal 508. In either case, there is a beneficial netreduction in effective load capacitance for the driver amplifier 30 andso the bandwidth and slew rate is extended.

As shown in FIG. 5, the heat sink may also be grounded (providing asafety advantage) with the positive feedback applied to a screeningconductor 606 via a direct connection (DC coupling terminal 602) or byAC coupling (capacitor 610 and AC coupling terminal 608) to the emitterof transistor Q4, the emitter of transistor Q7 or to the output terminal15. This approach is less thermally and electrically efficient but hascertain safety advantages such as avoiding a dangerous potential on theheat sink.

The examples of AC coupling of the feedback voltage to the heat sink orto the screen advantageously reduces the safety hazard at the heat sinkwhile remaining nearly as effective as DC coupling via terminals 502 or602.

What is claimed is:
 1. A display driver comprising: a video amplifiercoupled to a source of a video signal; an isolation means comprisingfirst and second transistors and having an input coupled to said videoamplifier and an output coupled to a kinescope cathode; first and secondfeedback circuits respectively coupled to said first and secondtransistors; and a current sensing circuit coupled to said isolationmeans for sensing current of said kinescope cathode during an inactiveportion of said video signal, said current sensing circuit coupled toautomatic kinescope bias circuitry, characterized by a coupling anddecoupling circuit disposed between said first transistor and saidoutput of said isolation means for coupling said first transistor tosaid output during an active portion of the video signal and decouplingsaid first transistor from said output during said inactive portion ofsaid video signal.
 2. The display driver according to claim 1,characterized in that said isolation means is a complimentary emitterfollower amplifier.
 3. The display driver according to claim 1,characterized in that said coupling and decoupling circuit includesthreshold conduction switch means coupled to an emitter of said firsttransistor and to said output of said isolation means.
 4. The displaydriver according to claim 3, characterized in that said first feedbackcircuit is coupled to a junction of said threshold conduction switchmeans and to said emitter of said first transistor.
 5. The displaydriver according to claim 3, characterized in that a capacitor iscoupled in parallel with said threshold conduction switch means betweensaid emitter of said first transistor and said output of said isolationmeans.
 6. The display driver according to claim 3, characterized in thatsaid first and second transistors are arranged in a push-pullconfiguration with respect to said output of said isolation means. 7.The display driver according to claim 3, characterized in that saidfirst transistor is of a first conduction type, said second transistoris of a second conduction type opposite to said first transistor, andsaid first and second transistors are arranged in a complimentarypush-pull configuration with respect to said output of said isolationmeans.
 8. The display driver according to claim 7, characterized in thatsaid first and second transistors are bi-polar transistors and saidrespective output electrodes are emitter electrodes.
 9. The displaydriver according to claim 7 or 8, characterized in that said thresholdconduction switch means is a diode.
 10. The display driver according toclaim 2, characterized in that said active portion of said video signalincludes video program information, said inactive portion of said videosignal corresponds to a blanking interval, and said current sensingcircuit is coupled to said second transistor for sensing current of saidkinescope cathode during said blanking interval.